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United States Patent6317452
Durrant , ; et al.November 13, 2001
Title
Method and apparatus for wireless spread spectrum communication with preamble sounding gap
Abstract
A wireless spread spectrum communication system comprises a spread spectrum transmitter and spread spectrum receiver which communicate according to an over-the-air protocol. The spread spectrum transmitter transmits a burst comprising a preamble followed by a short gap, followed by a data message. The spread spectrum receiver receives and demodulates the transmitted burst. The receiver detects the preamble using a non-coherent parallel correlator, and from the preamble correlation peak generates a series of integration periods for serial non-coherent correlation. The short gap between the preamble and the data message allows the receiver time to process the preamble and set the timing of the electronics for receiving the data message. The receiver has a plurality of non-coherent correlators operating in parallel to recover the spread spectrum encoded information. For each of M spread spectrum codes, the receiver simultaneously attempts to correlate the non-preamble portion of the received spread spectrum signal by separating the received signal into real and imaginary parts, correlating both real and imaginary parts for I and Q sequences, and combining the real I, real Q, imaginary I, and imaginary Q correlation signals into a unified correlation signal.
Inventors:Durrant; Randolph L. (Colorado Springs, CO), Burbach; Mark  (Peyton, CO)
Assignee:Xircom, Inc. (Thousand Oaks, CA)
Appl. No.:307646
Filed:May 7, 1999
Current U.S. Class:375/130 375/141 375/145 
Field of Search:375/141,142,146,147,150,130,133,145,149 370/320,335,342
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Primary Examiner: Tse; Young T.
Attorney, Agent or Firm:Lyon & Lyon LLP
Parent Case Text


This application is a continuation of application Ser. No. 08/928,846, filed on Sep. 12, 1997 now U.S. Pat. No. 5,953,370, which is a continuation of application Ser. No. 08/480,903, filed on Jun. 7, 1995 now abandoned, which is a continuation-in-part of application Ser. No. 08/304,091, filed on Sep. 9, 1994, now U.S. Pat. No. 5,648,982.
Claims

We claim:
1. A method for spread-spectrum communication between a first transceiver and a second transceiver, the method comprising the steps of:
transmitting, from a spread spectrum transmitter associated with the first transceiver to a spread spectrum receiver associated with the second transceiver, a spread spectrum preamble;
transmitting, from said spread spectrum transmitter to said spread spectrum receiver, a spread spectrum encoded data message immediately following said spread spectrum preamble, said spread spectrum encoded data message separated in time from said spread spectrum preamble by a predefined gap during which the second transceiver does not transmit to the first transceiver;
receiving said spread spectrum preamble at said spread receiver;
generating, at said spread spectrum receiver, a correlation signal in response to said spread spectrum preamble;
receiving said spectrum encoded data message at said spread spectrum receiver; and
despreading said spread spectrum encoded data message at said spread spectrum receiver.

2. The method of claim 1, wherein said predefined gap is of sufficient duration to allow correlation of said spread spectrum preamble by said spread spectrum receiver.

3. The method of claim 1, further comprising the steps of
generating a repeating time frame, and
defining a plurality of equally spaced time offsets relative to a starting boundary of said repeating time frame,
wherein said step of transmitting said spread spectrum preamble comprises the step of transmitting said spread spectrum preamble with reference to one of said time offsets.

4. A spread spectrum communication system, comprising:
a first transceiver having a first spread spectrum transmitter and a first spread spectrum receiver, said first spread spectrum transmitter transmitting a burst comprising a spread spectrum preamble and a spread spectrum encoded data message following said spread spectrum preamble after a preamble sounding gap; and
a second transceiver having a second spread spectrum transmitter and a second spread spectrum receiver, said second spread spectrum receiver comprising circuitry for receiving and correlating to said spread spectrum preamble and circuitry for receiving and processing said spread spectrum encoded data message;
wherein said second spread spectrum transmitter does not transmit to said first spread spectrum receiver during said preamble sounding gap.

5. A spread spectrum communication system, comprising:
a first transceiver comprising a spread spectrum transmitter; and
a second transceiver comprising a spread spectrum receiver;
wherein said spread spectrum transmitter and said spread spectrum receiver communicate using an over-the-air protocol according to which said spread spectrum transmitter transmits, and said spread spectrum receiver receives, a spread spectrum preamble followed after an idle period of predetermined duration by a spread spectrum encoded data message, said idle period being relatively short in comparison to a period of transmitting said spread spectrum encoded data message, no information being transmitted by said spread spectrum transmitter or received by said first transceiver during said idle period.

6. The spread spectrum communication system of claim 5, wherein said spread spectrum transmitter transmits a non-informational fill code during said idle period.

7. A multiple-access spread spectrum communication system, comprising:
a base station, said base station comprising a base station transmitter and a base station receiver; and
a plurality of user stations, each of said user stations comprising a user station transmitter and a user station receiver;
wherein each of said user stations transmits on an assigned channel a burst with reference to one of a plurality of time slots of a repeating time frame during which burst the user station does not receive information from the base station on the same channel, each burst comprising a spread spectrum preamble, a preamble sounding gap following said spread spectrum preamble, and a spread spectrum encoded data message following said spread spectrum preamble, said preamble sounding gap being shorter in duration than said spread spectrum encoded data message; and
wherein, for each burst, said base station receives said spread spectrum preamble, correlates said spread spectrum preamble during said preamble sounding gap, and despreads said spread spectrum encoded data message thereafter based upon the correlation of said spread spectrum preamble.

8. The multiple-access spread spectrum communication system of claim 7, wherein said spread spectrum preamble is the same for each of said user stations communicating with said base station.

9. The multiple-access spread spectrum communication system of claim 8, wherein said base station is located in a cell within a cellular communication system, wherein said spread spectrum preamble comprises a spread spectrum code selected from a set of spread spectrum codes available to each user station, and wherein each user station transmits a second spread spectrum preamble to a second base station located in a different cell from said base station, said second spread spectrum preamble comprising a different spread spectrum code than said first spread spectrum preamble.

10. The multiple-access spread spectrum communication system of claim 7, wherein said spread spectrum encoded data message comprises a sequence of M-ary spread spectrum codes of predefined length, and wherein said base station receiver comprises
means for dividing a received signal into a real signal and an imaginary signal;
a non-coherent parallel correlator for correlating to said spread spectrum preamble, said non-coherent parallel correlator receiving as inputs said real signal and said imaginary signal and generating, during the preamble sounding gap following receipt of said spread spectrum preamble, a correlation signal of variable width upon recognizing said spread spectrum preamble;
a center-seeking circuit for identifying the center of correlation signal and synchronizing a correlation clock thereby during said preamble sounding gap;
a bank of serial correlators, each of said serial correlators configured to recognize a different one of said M-ary spread spectrum codes and generate a correlation signal upon recognizing its respective M-ary spread spectrum code, said serial correlators each comprising an integrator responsive to said correlation clock and each receiving as inputs said real signal and said imaginary signal; and
a best-of-M magnitude comparator and data extractor connected to the output correlation signal from each of said serial correlators.

11. A method for multiple-access spread spectrum communication, comprising the steps of:
transmitting, from each of a plurality of user stations, a spread spectrum preamble;
transmitting, from each of said user stations, a spread spectrum encoded data message following the respective spread spectrum preamble transmitted by a user station, said spread spectrum encoded data message lagging said spread spectrum preamble by a preamble sounding gap during which the user station does not receive a data message, said preamble sounding gap shorter in duration than said spread spectrum preamble;
receiving said spread spectrum preambles at a receiver;
generating a correlation signal at said receiver in response to each spread spectrum preamble received at said receiver;
receiving said spread spectrum encoded data messages at said receiver, each spread spectrum encoded data message received following arrival of its respective spread spectrum preamble;
despreading each spread spectrum encoded data message at said receiver.

12. The method of claim 11, further comprising the steps of
generating, a plurality of time frames, and
defining a plurality of equally spaced time offsets relative to a starting boundary of each of said time frames,
wherein said step of transmitting, from each of said plurality of user stations, said spread spectrum preamble comprises the step of transmitting, from each of said plurality of user stations, said spread spectrum preamble with reference to one of said time offsets.
Description
FIELD OF THE INVENTION

The field of this invention relates to spread spectrum communication and, more particularly, to transmitting and receiving continuous phase modulated (CPM) signals such as spread spectrum signals.

DESCRIPTION OF THE RELATED ART

Spread spectrum is a type of signal modulation that spreads a signal to be transmitted over a bandwidth that substantially exceeds the data-transfer rate, hence the term "spread spectrum". In direct sequence spread spectrum, a data signal is modulated with a pseudo-random chip sequence; the encoded spread spectrum signal is transmitted to the receiver which despreads the signal. Several techniques are available for the transmitter to modulate the data signal, including biphase shift keying (BPSK) and continuous phase modulated (CPM) techniques. Minimum shift keying (MSK) is a known variation of CPM.

In despreading a spread spectrum signal, the receiver produces a correlation pulse in response to the received spread spectrum signal when the received spread spectrum signal matches the chip sequence to a predetermined degree. Various techniques are available for correlating the received signal with the chip sequence, including those using surface acoustic wave (SAW) correlators, tapped delay line (TDL) correlators, serial correlators, and others.

In spread spectrum communication CPM techniques are often chosen so as to preserve signal bandwidth of the spread spectrum signal when it is amplified and transmitted. Using CPM techniques also has the advantage that "class C" amplifiers may be used for transmitting the spread spectrum signal. However, spread spectrum signals transmitted using CPM are difficult to decode with many types of spread spectrum correlators, including various SAW correlators and serial correlators. These types of correlators usually require a BPSK spread spectrum signal for effective correlation rather than an MSK or other CPM spread spectrum signal because a BPSK signal has either a zero or 180 degree phase shift for each chip time. Thus, each chip of a received BPSK signal may be compared with each chip of the spread spectrum code, and a maximum correlation pulse may be generated when a predetermined number of matches occur. However, when a CPM signal with the same data signal and chip rate is applied to the same correlator, the correlation pulse will generally be very weak and may be quite difficult to detect.

Another problem often encountered in attempting to correlate spread spectrum signals transmitted using CPM techniques is the absence of a coherent reference signal in the receiver. A coherent reference signal in this sense may be defined as a locally generated signal that matches the transmitter carrier signal in frequency and phase. The receiver may use the locally generated reference signal to demodulate the received signal. In practice, however, it can be difficult to independently generate a local reference signal in the receiver precisely matching the transmitted carrier signal in frequency and phase. Rather, a local reference signal generated in the receiver will usually be of a non-coherent variety--that is, having small differences in frequency and phase from the transmitter's carrier signal. These frequency and phase differences are not constant but vary over time. When an attempt is made to demodulate a received signal using a non-coherent reference signal, errors in correlation may occur due to mismatches in timing and variations in perceived amplitude caused by the frequency and phase differences.

Various methods for dealing with the above problem exist in which a coherent reference signal is created in the receiver by continuously measuring the frequency and phase differences between the received signal and a locally generated non-coherent reference signal, and then adjusting the non-coherent reference signal until it matches the frequency and phase of the received signal. Such methods, however, generally require the use of relatively complex feedback techniques and involve extra hardware. Moreover, locking onto the received frequency and phase can take an unacceptably large amount of time, particularly in systems where time is of the essence, such as in certain time division multiple access (TDMA) systems in which only a relatively brief time slot is allocated for periodic communication between a transmitter and receiver.

A particular non-coherent digital matched filter is described in A. Baier and P. W. Baier, "Digital Matched Filtering of Arbitrary Spread-Spectrum Waveforms Using Correlators with Binary Quantization," 2 Proceedings, 1983 IEEE Military Communications Conference, Vol. 2, pp. 418-423 (1983). The digital filter described therein uses four real filter channels to perform four-phase quantization in the complex plane, with the four quadrants being the quantization regions, and the result taking on the four complex values of .+-.1.+-.j. In the described four-phase filter, an input signal is divided into an in-phase signal and a quadrature signal. The in-phase signal and the quadrature signal are separately filtered, sampled and digitized using 1-bit quantization. The quantized in-phase signal and the quantized quadrature signal are each fed into two binary correlators each programmed with a reference sequence of N chips, one chip for each sample. The outputs of the four binary correlators are combined to produce a resultant output signal. Baier's four-phase digital matched filter is also described in A. Baier, "A Low-Cost Digital Matched Filter for Arbitrary Constant-Envelope Spread Spectrum Waveforms," IEEE Transactions on Communications, Vol. Com-32, No. 4, April 1984, pp. 354-361.

These references suggest that for demodulation of non-coherent CPM signals such as QPSK, MSK, OQPSK, and GMSK signals, four real channels are needed to fully recover the transmitted signal. Further, the described four-phase filter shows only a system using 1-bit quantization, and does not describe a technique for serial correlation.

Accordingly, it would be advantageous to provide a method of modulation and demodulation particularly suited to CPM signals. It would further be advantageous to provide a method of CPM modulation and demodulation that does not require the generation of a coherent reference signal, that is capable of rapid correlation, and that may be used with analog correlators and digital correlators in an effective manner. It would further be advantageous to provide a flexible and effective system for CPM modulation and demodulation that does not require a coherent reference signal, and that is suitable for use in an environment of cellular communications.

SUMMARY OF THE INVENTION

The invention relates to a method and apparatus for transmitting and receiving CPM spread spectrum signals using phase encoding to increase throughput. In one aspect of the invention, a transmitter divides a signal data stream into a plurality of data streams (e.g., an I and Q data stream), independently modulates the data streams using CPM or a related modulation technique, and superposes the plurality of resultants for transmission. A preferred receiver receives the superposed spread spectrum signal, simultaneously attempts to correlate for a plurality of chip sequences (such as I and Q chip sequences), and interleaves the correlated data streams into a unified signal data stream.

In a second aspect of the invention, the receiver comprises a carrier signal that is neither frequency matched or phase matched with the transmitted signal. In this aspect, the receiver separates the received spread spectrum signal into real and imaginary parts, attempts to correlate both real and imaginary parts for a plurality of chip sequences (e.g., I and Q chip sequences), and combines the real and imaginary signals into a unified signal data stream. A preferred embodiment of this aspect of the invention uses a single bit digitization of the received spread spectrum signal to preserve only phase information for inexpensive digital processing. Another preferred embodiment of this aspect of the invention uses two-bit digitization of the received spread spectrum signal. In an alternative embodiment of the invention, the receiver uses self-synchronization techniques for despreading and correlation.

These aspects of the invention are described with reference to a preferred embodiment of the invention, in which a single parallel correlator and a plurality of 32 serial correlators are combined so as to allow correlation and recognition of any of 32 distinct symbols for a spread spectrum code sequence of 32 chips. Each of the 32 distinct symbols is associated with a distinct 5-bit pattern. A sixth bit of information is transmitted for each symbol by differential phase encoding at the transmitter and is phase decoded at the receiver.

A preferred transmitter capable of phase encoding divides a data stream into a data symbol portion and a phase selection portion. The data symbol portion is used to select one of a plurality of symbol codes for transmission. The phase selection portion is used to differentially phase encode the selected symbol code prior to transmission. The transmitter may use a CPM or related technique to transmit the phase encoded symbol codes.

A preferred receiver receives the superposed spread spectrum signal and simultaneously attempts to correlate for a plurality of chip sequences (such as I and Q chip sequences), and derives a real correlation signal and an imaginary correlation signal. For each received symbol, the receiver determines which is of a plurality of phase sectors the phase angle lies in. The receiver compares the difference between the phase sector of the present symbol and the phase sector of a preceding symbol. For biphase encoding, if the difference in closer to 0.degree., then the receiver outputs a first bit, and if the difference is closer to 180.degree., the receiver outputs a second bit. Higher degrees of phase encoding (e.g., quadraphase or octiphase) may also be used.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a spread spectrum communication transmitter and receiver as known in the art.

FIG. 2 depicts a pattern of cells for use in spread spectrum communication.

FIG. 3 is a graph of phase changes over time for an MSK signal.

FIGS. 4A-4C are a set of graphs showing a relationship among phase components.

FIG. 5A is a block diagram showing means for generating a CPM spread spectrum signal.

FIG. 5B is a graph of I and Q values.

FIG. 6 is a block diagram of a spread spectrum transmitter.

FIG. 7 is a block diagram showing one embodiment of spread spectrum receiver.

FIG. 8 is a block diagram showing another embodiment of a spread spectrum receiver.

FIG. 9 is a scatter diagram comparing transmitted and received I and Q signals.

FIG. 10 is a block diagram of an embodiment of a spread spectrum receiver using separable real and imaginary parts of a received spread spectrum signal.

FIGS. 11A-11F are diagrams showing a representation of transmitted and received waveforms for different phase values.

FIG. 12 is a block diagram of another embodiment of a spread spectrum receiver using separable real and imaginary parts of a received spread spectrum signal.

FIG. 13A is a block diagram of an embodiment of a spread spectrum receiver using serial correlation, and FIG. 13B is a waveform diagram associated therewith.

FIG. 14 is a block diagram of an embodiment of spread spectrum receiver using serial correlation for separable real and imaginary parts of the received spread spectrum signal.

FIG. 15A is a block diagram of another embodiment of a spread spectrum receiver using serial correlation for separable real and imaginary parts of the received spread spectrum signal.

FIG. 15B is a block diagram of a spread spectrum receiver using multi-bit serial correlation for separable real and imaginary parts of the received spread spectrum signal.

FIG. 15C is a graph showing an example of quantization of an I or Q waveform in accordance with the FIG. 15B receiver.

FIG. 15D is a block diagram of another embodiment of a spread spectrum receiver using multi-bit serial correlation for separable real and imaginary parts of the received spread spectrum signal.

FIG. 16 is a block diagram of an embodiment of spread spectrum receiver using self-synchronized correlation for separable real and imaginary parts of the received spread spectrum signal.

FIGS. 17A and 17D are block diagrams of a preferred transmitter and a preferred transmission protocol, respectively.

FIG. 17B is a diagram of an alternative transmission protocol.

FIG. 17C is an exemplary SQAM waveform generated by a transmitter using separate I and Q components.

FIG. 18 is a block diagram of a preferred noncoherent matched filter and associated receiver components.

FIG. 19 is a block diagram of a preferred digital circuit embodiment of a set of noncoherent serial correlators and associated receiver components.

FIG. 20 is a diagram showing exemplary correlation pulses within a predetermined timing window.

FIGS. 21A and 21B are schematic diagrams showing a preferred digital circuit embodiment of part of a receiving system used in conjunction with the circuitry of FIGS. 18 and 19.

FIG. 22 is a block diagram of a Robertson device for computing a sum of the squares of its inputs.

FIG. 23 is a block diagram of a correlator matched to a specific code sequence.

FIGS. 24A and 24B are digital circuit block diagrams of a spread spectrum transmitter employing differential phase encoding, and

FIG. 24C is a general block diagram thereof.

FIG. 24D is a diagram of an exemplary input data sequence and phase encoded symbol code output sequence.

FIGS. 25A and 25B-25C are block diagrams of two different embodiments of a receiver for carrying out phase decoding to obtain extra information from the received signal.

FIG. 26 is a block diagram of a preferred receiver for carrying out phase decoding in a 32 symbol transmission technique in accordance with the embodiment of the receiver shown in FIGS. 25B and 25C.

FIGS. 27A and 27B are phase map diagrams for an 8-sector phase map and a 16-sector phase map, respectively, and

FIG. 27C is a preferred 16-sector phase map diagram having a phase reference offset from zero.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 is a block diagram of a spread spectrum communication transmitter 101 and receiver 108 as known in the art.

The spread spectrum transmitter 101 of FIG. 1 comprises an input port 102 for input data 103, a transmitter chip sequence generator 104, and a modulator 105. The transmitter 101 thereby transmits a spread spectrum signal 106 over a transmission channel 107. The transmission channel 107 may comprise an RF channel, but may also comprise other transmission media, such as modulated laser, ultrasound, or fluidic systems. The spread spectrum receiver 108 of FIG. 1 comprises a receiver chip sequence generator 110, a demodulator 111, and an output port 112 for generating output data 113. In the FIG. 1 system, a single chip sequence, which appears essentially random to others not knowing the spreading code upon which it is based, may be identically generated by both the transmitter generator 104 and the receiver generator 110. An extensive discussion of spread spectrum communication, spreading codes, and chip sequences may be found in R. Dixon, Spread Spectrum Systems with Commercial Applications (J. Wiley & Sons, 3d ed. 1994).

FIG. 2 depicts a pattern of cells for use in spread spectrum communication.

In the preferred cellular environment of FIG. 2, a region 151 for communication may be divided into a set of cells 152, each of which may be assigned a frequency and a set of spread spectrum codes for communication. A first cell 153 may generally be found adjacent to a set of distance-one neighbors 154 and a set of distance-two neighbors 155. In a preferred embodiment, a plurality of frequencies f1, f2 and f3, and a plurality of code sets c1, c2, and c3, may be configured in a pattern of cells 152 so that the no distance-one neighbors 154 or distance-two neighbors 155 of a particular cell 153 has the same combination of frequency and code set as the cell 153.

Other and further information about a preferred cellular environment in which the invention may operate may be found in U.S. Pat. No. 5,402,413 which is assigned to the assignee of the present application, and hereby incorporated by reference as if fully set forth herein.

Known CPM spread spectrum signals include several variations; these include minimum shift keying (MSK) and its variations, e.g., Gaussian pre-filtered MSK (GMSK), superposed quadrature amplitude modulation (SQAM), and staggered quadrature offset raised cosine modulation (SQORC). These variations are known in the art. Explanations of various types of CPM techniques may be found in the following: Frank Amoroso and James A. Kivett, "Simplified MSK Signaling Technique," IEEE Transactions on Communications, April 1977, pp. 433-441; Mark C. Austin and Ming U. Chang, "Quadrature Overlapped Raised-Cosine Modulation," IEEE Transactions on Communications, Vol. Com-29, No. 3, March 1981, pp. 237-249; Kazuaki Murota and Kenkichi Hirade, "GMSK Modulation for Digital Mobile Radio Telephony," IEEE Transactions on communications, Vol. Com-29, No. 7, July 1981, pp. 1044-1050; and J. S. Seo and K. Feher, "SQAM: A New Superposed QAM Modem Technique," IEEE Transactions on Communications, Vol. Com-33, March 1985, pp. 296-300. The invention is generally described with regard to MSK signals. However, other variants of MSK and other CPM signals are within the scope and spirit of the invention.

An MSK signal is generally characterized by the fact that phase changes linearly within each chip time, and that the phase change over a single chip time is .+-..pi./2 radians (.+-.90 degrees). The rate of phase change for a single chip time is .+-.k, for a suitable value k, and is linear and continuous everywhere except at chip boundaries.

The above described characteristics of MSK signals may be further explained with reference to FIG. 3, which is a graph showing possible changes in phase for an MSK signal over time. In FIG. 3, the x-axis is time and the y-axis is signal phase. In a first chip time from zero to Tc, the phase .theta.(t) changes from 0 to .pi./2 or -.pi./2. In a second chip time, from Tc to 2 Tc, the phase .theta.(t) changes from +.pi./2 to 0 or +.pi./2 to +.pi., or from -.pi./2 to 0 or -.pi./2 to -.pi., and so on.

An MSK signal s(t) may be considered to comprise two offset signals, i(t) and q(t), which represent the phase of the carrier signal. At any instant of time the phase of the carrier signal may be expressed as:

Thus, i(t)=cos .theta.(t), and q(t)=sin .theta.(t).

Since the phase of the MSK signal varies linearly from one chip time to the next chip time, i(t) and q(t) may consist of half sinusoidal waveforms as shown in the FIGS. 4A-4C. In FIGS. 4A-4C, the x-axis is time and the y-axis is signal phase. FIG. 4A is a graph showing an example of how the phase .theta.(t) may change for a particular MSK signal in each chip time from 0, Tc, 2Tc, 3Tc, 4Tc, 5Tc, and so on, for the chip sequence "11101001 . . . ." As noted, during each chip time the phase varies for an MSK signal by .pi./2 in either a positive or negative direction. FIGS. 4B and 4C are graphs showing i(t) and q(t) waveforms, respectively, which correspond to the varying phase .theta.(t). Because of the nature of the MSK signal's phase .theta.(t) (i.e., that it is linear and varies only by .pi./2 each chip period), the i(t) signal comprises a sequence of partial cosine waveforms, and the q(t) signal comprises a sequence of partial sine waveforms. Each of i(t) and q(t) comprises a half-waveform over a timespan of 2Tc; that is, i(t) and q(t) occur at half the chip rate.

An i(t) waveform and a q(t) waveform can be generated from a chip stream c(t) and combined so as to produce an MSK signal--i.e., a signal having a phase which varies linearly as desired in either a positive or a negative direction by an amount of .pi./2 each chip time. In order to generate i(t) and q(t), the original chip stream c(t) may be demultiplexed into two separate chip streams C.sub.even (t) and C.sub.odd (t), each having half the chip rate of the original chip stream c(t). In the described embodiment, the i(t) signal is associated with the odd-numbered chips, and the q(t) signal is associated with the even-numbered chips.

Thus, the i(t) signal comprises a sequence of half-sinusoidal waveforms, one for each odd chip. Each half sinusoid may be positive for a "1" chip and negative for a "0" chip:

where C.sub.odd (t) comprises the odd-numbered chips from the chip stream to be transmitted. Similarly, the q(t) signal comprises a sequence of half-sinusoidal waveforms, one for each even chip:

where C.sub.even (t) comprises the even-numbered chips from the chip stream to be transmitted.

The i(t) and q(t) signals may be used to modulate a carrier signal operating at frequency .omega..sub.0 by summing i(t) and q(t) in phase quadrature so as to generate an MSK signal s(t) having a linearly varying phase .theta.(t). A block diagram showing means for generating a CPM spread spectrum signal is depicted in FIG. 5A. The signal i(t) is multiplied with a signal A cos .omega..sub.0 t by multiplier 250, which provides an output to a summer 252. The signal q(t) is multiplied with a signal A sin .omega..sub.0 t by multiplier 251, which also provides an output to the summer 252. The summer 252 sums its inputs and produces an output signal s(t).

The relationship between the transmitted signal s(t) having varying phase .theta.(t), and the i(t) and q(t) signals is shown in the following equations:

s(t) = Re {A exp(j [-.omega..sub.0 t + .theta.(t)])} = Re {A exp(-j.omega..sub.0 t) exp(j.theta.(t))} = Re {A [cos .omega..sub.0 t - j sin .omega..sub.0 t] [i(t) + jq(t)]} = A i(t) cos .omega..sub.0 t + A q(t) sin .omega..sub.0 t (207)

where A is an amplification factor, Re{} represents the real part of a complex value, and j is the square root of -1. Note that u(t)=i(t)+jq(t) represents the complex envelope of s(t).

As noted herein, i(t) and q(t) each comprises every other chip from the chip stream c(t); i(t) comprises the odd-numbered chips 1, 3, 5, . . . ; q(t) comprises the even-numbered chips 2, 4, 6, . . . . The transmitted signal s(t), generated from signals i(t) and q(t), therefore comprises all of the chips. Because q(t) is derived from the even chips while i(t) is derived from the odd chips, q(t) is delayed by one chip time from i(t); thus, q(t) and i(t) are offset signals.

It is important to note that, because i(t) and q(t) are staggered, as i(t) reaches its maximum (or minimum) value q(t) will be zero, and vice versa. This relationship between i(t) and q(t) allows phase change sequences of .+-..pi./2 over one chip time Tc (unlike, for example, QPSK or OQPSK). FIG. 5B is a graph of I and Q values, in which the x-axis represents values of i(t) and the y-axis represents values of q(t). Each <i(t), q(t)> pair falls at a given instant of time on the circle
260. Maximum and minimum values for i(t) and q(t) are shown where the circle 260 intersects the x-axis and y-axis at points 265 through 268; these points 265 through 268 also represent the possible values of <i(t), q(t)> pairs at chip boundary times.

Alternative encoding methods such as GMSK, SQAM, or SQORC, differ from MSK in that phase changes of less than .+-..pi./2 are allowed. In general, GMSK, SQAM, and SQORC all use a form of pre-filtering the MSK i(t) and q(t) signals to reduce transmission bandwidth. This pre-filtering has the general effect of reducing the high-frequency components generated by the sharp phase reversals in the MSK i(t) and q(t) signals. For GMSK, pre-filtering may also result in intersymbol interference over several chip times, the effect of which may be mitigated with a trellis decoder. In SQAM or SQORC, the final frequency envelope is no longer constant, but is still nearly so.

FIG. 6 is a block diagram of a spread spectrum transmitter.

In the transmitter of FIG. 6, a chip stream c(t) 301 is provided to a demultiplexer 302, which divides the chip stream 301 into a set of odd chips C.sub.odd (t) 303 for the i(t) signal and a set of even chips C.sub.even (t) 304 for the q(t) signal. The chip stream c(t) 301 may comprise the result of a pseudo-noise ("PN") code modulated with a data stream (as in direct sequence spread spectrum communication), or may comprise a sequence of chip codes corresponding to predetermined symbols such as may be done, for example, in code shift keying (CSK) techniques.

The odd chips 303 and the even chips 304 are each coupled to first and second waveform generators P(t) 305 and 306 respectively. In a preferred embodiment, the waveform generators P(t) may generate a half-sinusoidal waveform, positive or negative, as described herein. Other waveform generators and other waveforms are within the scope and spirit of the invention.

The output of the first waveform generator 305 (i.e., receiving the odd chips 303) corresponds to the signal i(t) and is coupled to a first multiplier 307, which modulates a carrier signal cos .omega..sub.0 t to generate a signal s.sub.1 (t) 308
corresponding to i(t) cos .omega..sub.0 t. The output of the second waveform generator 306 (i.e., receiving the even chips 304) corresponds to the signal q(t), which, as mentioned, is delayed by one chip time Tc from the signal i(t). The output of the second waveform generator 306 is coupled to a second multiplier 310, which modulates a carrier signal sin w.sub.0 t to generate a signal s.sub.2 (t) 311 corresponding to q(t) sin w.sub.0 t.

The signals s.sub.1 (t) 308 and s.sub.2 (t) 311 are coupled to a summer 312, which combines its inputs and generates a superposed signal s(t) 313. The signal s(t) may be amplified and transmitted by a transmission system, such as a radio transmission system, coupled to the transmission channel 107.

The chip stream c(t) may be generated by modulating a pseudo-noise code with data to be transmitted such as is known in direct sequence spread spectrum modulation. In a preferred embodiment, the chip stream c(t) comprises a plurality of symbol codes, each symbol code representing a symbol indicative of one or more data bits of information. Instead of directly modulating input data with a pseudo-noise code, sequences of data bits are translated into symbols which are used to select from a plurality of symbol codes located in a table. For example, five data bits may represent a symbol; thus, there may be 32 possible symbols representing all possible combinations of five data bits. Each symbol is associated with a unique symbol code, so that thirty-two symbol codes (or sixteen symbol codes and their inverses) may represent all possible symbols. For each symbol to be transmitted, the appropriate symbol code is selected among the thirty-two available. Thus, the chip stream c(t) may comprise a sequence of symbol codes.

Each symbol code may be, for example, 32 chips in length, or some other appropriate number of chips in length (preferably an even number of chips).

In a like manner, the demultiplexer 302 may comprise a table of half symbol codes. In particular, the demultiplexer 302 may comprise a Q-lookup table and I-lookup table. For every five bits of data to be transmitted (following the previous example), instead of looking up a symbol code from a table and demultiplexing it with demultiplexer 302, two half symbol codes may be read, one from the I-lookup table and one from the Q-lookup table. Each half symbol code may be clocked serially to the waveform generators 305, 306 for further processing. The system may comprise clocking logic which provides a delay of one chip time Tc to the half symbol code from the Q-lookup table.

Once a set of 32 unique symbol codes are selected, the contents of the I-lookup table and Q-lookup table can be generated by dividing each symbol code into even and odd chips, and using the even chips for the half symbol codes in the Q-lookup table and the odd chips for the half symbol codes in the I-lookup table. Other techniques for generating even and odd chip sequences suitable for signals q(t) and i(t) fall within the spirit and scope of the invention.

FIG. 7 is a block diagram of a spread spectrum receiver.

The transmitted signal s(t) 313 may undergo attenuation, addition of noise, multipath superposition, and other known and unknown effects of the transmission channel 107. Accordingly, the received signal s*(t) 401 may differ from the transmitted signal s(t) 313 in known and unknown ways.

Received signal s*(t) may be despread using multiple correlators keyed to I and Q chip streams. Because CPM spread spectrum signals may be thought of as the superposition of time staggered signals created from I and Q chip streams (each at half the chip rate), a receiver according to one embodiment of the present invention uses two correlators, one programmed with the I-chip-sequence and one programmed with the Q-chip-sequence and both operating at half the chip rate, to decode the received signal, and then combines the outputs of the two correlators.

In the receiver of FIG. 7, the received signal s*(t) 401 is coupled to a CPM correlator 402 for recognizing a chip sequence in the received signal s*(t) 401. The CPM correlator 402 comprises a power divider 403 for generating duplicate signals, an i*(t) signal 404 with a 0 degree phase delay, and a q*(t) signal 405 with a 90 degree phase shift.

The i*(t) signal 404 is coupled to a delay 406, which delays the i*(t) signal 404 by one chip time Tc to allow simultaneous generation of correlation pulses by the I correlator 407 and the Q correlator 409. Thus, the delayed i*(t) signal is coupled to an I correlator 407, and the q*(t) signal 405 is coupled directly to a Q correlator 409.

The I correlator 407 operates at a chip rate of Rc/2, where Rc is the chip rate (i.e., 1/Tc) of the received signal s*(t) 401. The I correlator 407 may comprise one of several types of correlators, e.g., a surface-acoustical-wave (SAW) correlator, a tapped-delay-line (TDL) correlator, or a serial correlator. Examples of suitable correlators may be found in U.S. Pat. No. 5,016,255 entitled "Asymmetric Spread Spectrum Correlator" or in U.S. Pat. No. 5,022,047 entitled "Spread Spectrum Correlator", both of which are issued in the name of inventors Robert C. Dixon and Jeffrey S. Vanderpool and hereby incorporated by reference as if fully set forth herein. The I correlator 407 produces an output I correlation signal 408
indicating a degree of match between the delayed i*(t) signal and a predetermined I-chip-sequence.

The Q correlator 409 similarly operates at a chip rate of Rc/2, and may similarly comprise any of a number of suitable correlators such as those described in the patents referenced in the preceding paragraph. The Q correlator 409 produces an output Q correlation signal 410 indicating a degree of match between the q*(t) signal and a predetermined Q-chip-sequence.

The I correlation signal 408 and the Q correlation signal 410 are coupled to a summer 411, which combines its inputs and produces a unified correlation signal 412. Because the i*(t) signal is delayed by delay 406, the I correlation signal 408
and Q correlation signal 410 occur simultaneously. The unified correlation signal 412 is used to determine a data stream d(t) from which the chip sequence c(t) was generated.

The I correlator 407 and the Q correlator 409 thus jointly identify the chip sequence in the received signal s*(t) 401. The I correlator 407 is configured to recognize the odd chips of the chip sequence, while the Q correlator 409 is configured to recognize the even chips of the chip sequence. When the entire correlation sequence appears in the received signal s*(t), the sum of the I correlation signal 408 and the Q correlation signal 410 is at a maximum, and may be compared against a predetermined threshold to allow recognition of the chip sequence. A unified correlation signal 412 is produced when a chip sequence is recognized.

Alternatively, instead of comparing the unified correlation signal 412 to a predetermined threshold, a system may be configured so as to have a plurality (e.g., 32) of CPM correlators 402 operating in parallel, each tuned to recognize a different code sequence. The outputs of all 32 CPM correlators may be summed and, when the sum is at a predetermined maximum level, the CPM correlator 402 with the highest magnitude output may be chosen by a best-of-M detector or similar means as indicative of the data stream d(t). For example, in a CSK system, each of 32 CPM correlators may attempt in parallel to recognize a code sequence, and the one with the highest magnitude correlation signal may be assumed to indicate the received chip stream. The recognized chip stream may correspond to a data symbol from which a portion of the data stream d(t) may be recovered.

In a preferred embodiment, the CPM correlator 402 may be used in conjunction with techniques described in U.S. Pat. Nos. 5,016,255 or 5,022,047, both of which are assigned to the assignee of the present invention and hereby incorporated by reference. In those techniques, each data bit or data symbol of the data stream d(t) may be encoded by modulation with the entire length of a pseudo random chip sequence generated from a chip sequence code. For example, if a chip sequence code identifies a pseudo random chip sequence that repeats after 32 chips, each data bit of the data stream d(t) may be modulated with all 32 of those chips.

However, there is no requirement that the CPM correlator 402 be used with those particular techniques. For example, the CPM correlator may be used with other spread spectrum techniques to recognize a correlation signal that is used to synchronize the transmitter 101 and the receiver 108. Also, the CPM correlator 402 may be used in conjunction with a self-synchronizing or auto-synchronizing spread spectrum technique such as described elsewhere herein in more detail.

The I and Q chip sequences are preferably of equal length; thus, each CSK symbol code is preferably an even number of chips in length so as to avoid a 90-degree phase uncertainty between symbol codes when despreading is attempted.

FIG. 8 is a block diagram of a coherent spread spectrum receiver.

The received signal s*(t) 401 in the receiver of FIG. 8 is coupled to a CPM correlator 502 for recognizing a chip sequence in the received signal s*(t) 401. The CPM correlator 502 comprises a power divider 503, which produces duplicate signals
504 and 505, each with a 0 degree phase delay. Such power dividers are known in the art and are generally preferred for the CPM correlator 502 over the power divider 403 shown in FIG. 7. While a phase delay of 90 degrees between i*(t) and q*(t) was imposed by use of the power divider 403 in FIG. 7, a 90-degree phase delay in the FIG. 8 embodiment is produced by separately multiplying the signals 504 and 505 with cosine and sine signals, respectively.

The signal 504 is multiplied with a cos w.sub.0 t signal by I multiplier 530 and filtered by a I low pass filter 506 to provide an i*(t) signal. The signal 505 is multiplied by a sin .omega..sub.0 t signal by Q multiplier 531 and filtered by a Q low pass filter 512 to provide a q*(t) signal.

The outputs of the I low pass filter 506 and the Q low pass filter 512 generally appear for MSK as half sinusoidal waveforms corresponding to those generated in the transmitter from P(t) generators 305, 306.

The i*(t) signal output from I low pass filter 506 is coupled to an I correlator 507. The I correlator 507 comprises a register 508 having a sequence of chips 509. The register 508 may be an analog shift register, a tapped delay line having a plurality of taps, or any other suitable storage means. The odd chips are coupled by a plurality of multipliers to an I summer 510, which combines its inputs and produces an output I correlation signal 511.

An example of the path of the I correlator 507 is shown in FIG. 23. As described with respect to FIG. 8, the filtered i*(t) signal is coupled to a register 508. The register 508 comprises a series of chips 509 along which the filtered i*(t) signal propagates. The register 508 is matched to a particular code sequence. Thus, in the example of FIG. 23, the sequence of odd chips which will result in a match is C.sub.odd (t)=11001000. At time t=16Tc, the first chip C.sub.1 is compared with the first chip in the sequence of C.sub.odd (t), and a "1" is generated if the chips are equal. Each of the other odd chips in the register 508 is likewise compared against the programmed sequence. A comparison between any two chips may be carried out using a multiplier or an exclusive-OR gate. The comparison values are provided to a summer 510 which generates a maximum pulse when the chip sequence for which the correlator 507 has been programmed matches the received chip sequence. In FIG. 23, the branches having a "-1" correspond to chips for which a "0" in the received chip sequence will generate a match, while the other branches correspond to chips for which a "1" in the received chip sequence will generate a match.

Returning to FIG. 8, the q*(t) signal output from the Q low pass filter 512 is coupled to a Q correlator 513. The Q correlator 513 similarly comprises a register 514 having a sequence of chips 515. As with the odd chips in the I correlator 507, the even chips are coupled to a Q summer 516, which combines its inputs and produces an output Q correlation signal 517.

The I correlation signal 511 and the Q correlation signal 517 are coupled to a summer 518, which combines its inputs and produces a unified correlation signal 519. Because the I correlation signal 511 is derived from the odd chips while the Q correlation signal 517 is derived from the even chips (which precede the odd chips by one chip time Tc), the correlation signals 511, 517 occur simultaneously, and there is no need for a separate delay element such as delay 406 shown in FIG. 7. The unified correlation signal 519 is used to determine a data stream d(t) from which the chip sequence c(t) was generated in a manner similar to that explained above with reference to FIG. 7.

The FIG. 8 receiver operates best with a coherent carrier reference .omega..sub.0 and assumes such is available. Methods are known in the art for obtaining a coherent carrier reference, such as the use of phase estimating circuitry. Where very rapid acquisition times are necessary, such as in certain high-speed time division multiple access (TDMA) systems using CPM spread spectrum techniques, other embodiments (such as the non-coherent receiver embodiments described herein) may generally be preferred.

In a non-coherent CPM system, the receiver 108 of FIG. 1 may not have available an exact copy of the carrier signal at frequency .omega..sub.0 used by the transmitter 101. Rather, the receiver 108 generates a local carrier signal having a frequency .omega..sub.1, which in practice may differ in frequency and phase from the transmitter's carrier signal:

where .DELTA..omega.=frequency difference and .theta.=phase difference.

FIG. 10 is a block diagram of a non-coherent spread spectrum receiver for receiving and despreading a CPM spread spectrum signal without the need for a locally generated coherent reference signal .omega..sub.0. The receiver of FIG. 10 can be used to process a received CPM signal by splitting the received spread spectrum signal into separable real and imaginary parts, splitting the real and imaginary parts into I and Q portions, mixing the real I and Q portions and the imaginary I and Q portions with a non-coherent reference signal having a frequency near that expected of the received signal to obtain real I and Q streams and imaginary I and Q streams, filtering the multiplied signals, correlating separately the I and Q streams for each of the real and imaginary parts to obtain a real I and Q correlation pulse and an imaginary I and Q correlation pulse, combining the I and Q correlation pulses separately for the real and imaginary parts to provide a combined real and a combined imaginary correlation signal, squaring the combined real and imaginary correlation signals to generate a squared real and a squared imaginary correlation pulse, and combining the squared real and imaginary correlation signals into a unified correlation signal.

The operation of the receiver of FIG. 10 may be explained graphically with reference to FIG. 9, which is a scatter diagram comparing real and imaginary values as transmitted and as received in a non-coherent receiver. For simplicity, the explanation below assumes the transmission channel to be distortionless and have unlimited bandwidth. The transmitter's coordinate system 601 is represented by an x-axis and y-axis, with the x-axis representing values of i(t) and the y-axis representing values of q(t). A set of four points 610 through 613 represents transmitted sampled value pairs for <i(t.sub.n),q(t.sub.n)>. The pairs 610 through 613 represent coordinates <1,0>, <0,1>, <-1, 0>, and <0,-1>, respectively.

A receiver's coordinate system 604 is represented by an x*-axis and a y*-axis shown as dashed lines in FIG. 9. The receiver's coordinate system 604 is assumed to differ from the transmitter's coordinate system 601 due to frequency and phase differences. The receiver's coordinate system 604 rotates with respect to the transmitter's coordinate system 601 at a rate proportional to .DELTA..omega., the frequency difference ("beat frequency") between the transmitter and receiver reference signals. For sufficiently small .DELTA..omega. (such as may be expected for the time period of interest over which correlation for a data symbol will occur--e.g., 32 chip periods), the receiver's coordinate system 604 approximately equals the transmitter's coordinate system 601, except for a phase difference .theta. which remains relatively constant for short periods of time. In order to maintain such a condition, the beat frequency .DELTA..omega. preferably should be less than about 1/4
the symbol rate. For example, with a symbol rate of 156.25k symbols/second (5 Mchips/second), the beat frequency .DELTA..omega. should be less than about 39 kHz for optimal operation.

Because the receiver's coordinate system 604 at a given instant appears rotationally shifted with respect to the transmitter's coordinate system 601, the <i*(t.sub.n),q*(t.sub.n)> sampled pair recognized by the receiver 108 will be a point on the circle 607 corresponding to an <i(t.sub.n),q(t.sub.n)> sampled pair in the transmitter's coordinate system 601 but shifted around circle 607 by an amount dependent on the phase difference .theta.. Accordingly, the perceived real value or i*(t) will differ from the transmitted i(t) value by an amount dependent upon cos .theta. due to the rotational difference between the coordinate systems 601 and 604, while the perceived imaginary value or q*(t) will also differ from the transmitted q(t) value by an amount dependent upon sin .theta. for the same reason. Thus, if the transmitted <i(n), q(n)> sampled values are <1, 0> and the phase offset .theta. is +30.degree., the received <i*(t.sub.n), q*(t.sub.n)> sampled values are <cos +30.degree., sin +30.degree.> or <0.866, 0.5>. Likewise, if the phase offset .theta. is +90.degree. for the same transmitted values, the received <i*(t.sub.n), q*(t.sub.n)> sampled values are <0, 1>.

From the above explanation, it is apparent that a correlator attempting to correlate for I and Q portions would be faced with a diminishing i*(t) value as .theta. varies from 0 to 90 degrees, yet at the same time an increasing q*(t) value. As .theta. grows, eventually the difference between <i(t), q(t)> and <i*(t), q*(t)> becomes so large that accurate correlation is cumbersome. Because of the phase difference .theta., it is generally not possible to know in advance which quadrant of FIG. 9 the received signal s*(t) will be in relative to the transmitter's coordinate system 601. However, the present invention in one aspect addresses this problem by utilizing both real and imaginary parts of I and Q portions in order to despread the received s*(t) signal.

It may be noted that as the real portion of i*(t) decreases as .theta. varies from 0 to 90 degrees, the imaginary portion of i*(t) increases. Similarly, as the real portion of i*(t) increases (in magnitude) as .theta. varies from 90 to 180
degrees, the imaginary portion of i*(t) decreases. A similar phenomenon occurs with the real and imaginary portions of q*(t). The receiver of FIG. 10 takes advantage of the complementary aspects of the real and imaginary portions of the received i*(t) and q*(t) signal portions, and accordingly analyzes both the real and imaginary parts of the I and Q signals in order to make an effective correlation.

In the FIG. 10 embodiment, the received signal s*(t) 401 is coupled to a non-coherent CPM correlator 702 for recognizing a correlation sequence in the received signal s*(t) 401. The non-coherent CPM correlator 702 comprises a power divider 703, which produces duplicate signals Real*(t) 704 having a 0-degree phase delay and Imag*(t) 705 having a 90-degree phase delay. Real*(t) 704 and Imag*(t) 705 may be viewed as the real and imaginary parts of the received signal s*(t) 401.

The Real*(t) signal 704 is coupled to a CPM correlator 715 similar to CPM correlator 502 of FIG. 8, with the exception that the local reference signal is different, as described below. The CPM correlator 715 produces a real correlation signal
706. The Imag*(t) signal is coupled to a second CPM correlator 715 which produces an imaginary correlation signal 707. The real correlation signal 706 is coupled to a squaring device 708, which computes the square of its input. The imaginary correlation signal 707 is likewise coupled to a squaring device 709, which computes the square of its input. The outputs of the squaring devices 708 and 709 are coupled to a summer 710, which combines its inputs to produce a unified correlation signal
711 which is the sum of the squares of the real correlation signal 706 and the imaginary correlation signal 707. The unified correlation signal 711 is coupled to a square root device 712 which takes the square root of its input, and generates a final correlation signal 713 comprising correlation pulses 714. The time between correlation pulses 714 may be one symbol code time period Ts if CSK is employed.

A primary difference between the CPM correlators 715 shown in FIG. 10 and the CPM correlator 502 of FIG. 8 is that the CPM correlators 715 of FIG. 10 utilize non-coherent reference signals cos .omega..sub.1 t=cos(.omega..sub.0
+.DELTA..omega.)t+.theta. and sin .omega..sub.1 t=sin(.omega..sub.0 +.DELTA..omega.)t+.theta. for the I and Q portions, respectively, rather than cos .omega..sub.0 t and sin .omega..sub.0 t as generated in the coherent receiver of FIG. 8. The reference signals cos .omega..sub.1 t and sin .omega..sub.1 t may be generated from the same oscillator coupled to a power divider to keep the phase offset .theta. the same for both cos .omega..sub.1 t and sin .omega..sub.1 t. The use of non-coherent reference signals causes the correlation signal generated by each CPM correlator 715 to have a magnitude dependent in part upon the phase difference .theta..

The effect of using non-coherent reference signals on the ability to achieve correlation may be explained first with reference to the I portion of the Real*(t) signal 704. The Real*(t) signal 704 may be represented as:

where, as mentioned previously, u(t)=i(t)+jq(t), which is the complex envelope of s(t), and Re {} denotes the real portion of a complex value. The Real*(t) signal 704 is multiplied by multiplier 720 with a locally generated reference signal cos .omega..sub.1 t=cos(.omega..sub.0 +.DELTA..omega.)t+.theta., so that the output of multiplier 720 is:

The output of the multiplier 720 is coupled to a low pass filter 721 which retains the baseband portion of the signal coupled to its input. Assuming that the non-coherent reference signal cos .omega..sub.1 t differs from the transmitter reference frequency .omega..sub.0 by only a phase difference (i.e., that the frequency change is negligible over the time period of interest), then the receiver reference signal may be expressed as:

The output y(t) of the low pass filter 721 may therefore be expressed as:

y(t) = LPF [ Re {A u(t) exp(-j.omega..sub.0 t)} cos .omega..sub.1 (t) ] = LPF [ Re {A u(t) exp[j(-.omega..sub.0 t + .omega..sub.1 t)]} ] = (A/2) i(t) cos(.omega..sub.0 + .omega..sub.1 t)t + (A/2) q(t) sin(.omega..sub.0 t + .omega..sub.1 t) = (A/2) i(t) cos(-.theta.) + (A/2) q(t) sin (-.theta.) = (A/2) i(t) cos .theta. - (A/2) q(t) sin .theta. (790)

where "LPF" denotes operation of the low pass filter 721.

By similar deduction the output z(t) of the low pass filter 731 of the Q portion of the Real*(t) signal is as follows:

z(t) = (A/2) i(t) sin(-.theta.) + (A/2) q(t) cos(-.theta.) = (-A/2) i(t) sin .theta. + (A/2) q(t) cos .theta. (791)

Due to the 90-degree phase shift in signal 705, the output of low pass filter 741 of the I portion of the Imag*(t) signal is equal to z(t) as derived above, while the output of low pass filter 743 of the Q portion of the Imag*(t) signal is equal to the inverse of y(t) as derived above.

In operation, each of the four correlators 722 through 725 may contribute to correlation of the received CPM signal s*(t). Operation of the non-coherent CPM correlator 702 may be shown through several examples. As a first example, assume that the phase offset .theta.=0.degree.; therefore, the outputs y(t) and z(t) for low pass filters 721 and 731, respectively, reduce to the following:

and

Selecting an amplification factor A=2, the filter outputs of filters 721 and 731 then become y(t)=i(t) and z(t)=q(t). Assuming, for convenience, a code sequence length of 16 chips, then after 16 chip times (i.e., 16Tc) the entire sequence is contained within the correlation registers 726, 727, 728, and 729 in each CPM correlator 715. An illustrative chip stream c(t)=1111010110010000 may be broken into sub-sequences C.sub.odd (t)=11001000 and C.sub.even (t)=11110100. It will further be assumed for sake of explanation that the waveform generator P(t) of the transmitter generates a return-to-zero (RZ) rectangular waveform having a duration of two chip periods, so that the transmitted i(t) and q(t) signals may be depicted as shown in FIG.
11A and FIG. 11B, respectively. Operation of the FIG. 10 correlator using CPM baseband signals instead of RZ signals can be understood by observing that at time t=16Tc, the peak values of the sinusoidal waveforms appear in the correlation registers 726,
727, 728 and 729, and correspond to the pulse height of the RZ waveform.

At the receiving end, the contents of the correlation registers 726 and 727 may be represented as shown in FIGS. 11C and 11D, respectively. It can be seen that the waveform of FIG. 11C as reading from right to left is the same as that of FIG.
11A as reading from left to right. Similarly, the waveforms of FIGS. 11B and 11D bear the same relationship. An output for each of the four correlators 722, 723, 724 and 725 may be obtained by pointwise multiplication of the chip values with the chip weighting factors 716 for each chip, and summation of the chip products by summers 717 to produce a correlation signal. The chip weighting factors 716 for correlator 725 are opposite in sign to the values for correlator 723. The chip weighting factors
716 for correlators 722 and 724 are the same sign.

Continuing with the present example in which .theta.=0.degree., the output at time t=16Tc for each of correlators 722 and 723, corresponding respectively to the I portion ("ReI") and the Q portion ("ReQ") of the Real*(t) signal, is eight, while the output for each of correlators 724 and 725, corresponding respectively to the I portion ("ImI") and the Q portion ("ImQ") of the Imag*(t) signal, is 0. The final correlation signal 713 at the instant 16Tc is: ##EQU1##

The value of 16 is a maximum value indicating correlation for the particular chip sequence. If multiple codes are to be recognized, a plurality of non-coherent CPM correlators 702 may operate in parallel, each programmed to recognize a different code. The chip sequence corresponding to the highest correlation signal may be selected as the received chip sequence.

Assuming as a second example that .theta.=30.degree., the contents of correlation registers 726 and 727 appear as shown in FIGS. 11E and 11F, respectively. Selecting the amplification factor A=2, the outputs y(t) and z(t) of low pass filters 721
and 731, respectively, may be represented as: ##EQU2##

Pointwise vector multiplication of each of the chip valves in the correlation registers 726 through 729 with corresponding chip weights 716 yields the following outputs from summers 717:

ReI = (1)(0.866) + (1)(0.866) + (-1)(-0.866) + (-1)(-0.866) . . . = (8)(0.866) = 6.928 ReQ = (1)(0.866) + (1)(0.866) + (1)(0.866) . . . = (8)(0.866) = 6.928 ImI = (1)(-0.5) + (1)(-0.5) + (-1)(0.5) + (-1)(0.5) . . . = -(8)(0.5) = -4.0 ImQ = (1)(-0.5) + (1)(-0.5) + (1)(-0.5) + (1)(-0.5) . . . = -(8)(0.5) = -4.0

A final correlation signal 713 therefore is generated:

Thus, for a phase offset of .theta.=30.degree., the value of the final correlation signal 713 at t=16Tc remains at the maximum level of 16.

As another example, a phase offset .theta.=45.degree. is assumed. The outputs y(t) and z(t) of low pass filters 721 and 731, respectively, become:

and

Solving for the intermediate values ReI, ReQ, ImI, and ImQ yields:

ReI = (1)(0.707) + (1)(0.707) . . . = (8)(0.707) = 5.657 ReQ = (1)(0.707) + (1)(0.707) . . . = (8)(0.707) = 5.657 ImI = (1)(-0.707) + (1)(-0.707) . . . = -(8)(0.707) = -5.657 ImQ = (1)(-0.707) + (1)(-0.707) . . . = -(8)(0.707) = -5.657

A final correlator signal 713 is generated:

Again, maximum correlation of 16 is realized even though the phase offset .theta. is not equal to 0.

A table can be constructed of (ReI+ReQ), (ImI+ImQ) values and correlation values versus phase offset .theta. for the correlator of FIG. 10:

.theta. R.sub.i + R.sub.q I.sub.i + I.sub.q Corr= 0.degree. 16 0.0 16.0 30 13.856 -8.0 16.0 45 11.314 -11.314 16.0 60 8.0 -13.856 16.0 90 0.0 -16.0 16.0 120 -8.0 -13.856 16.0 135 -11.314 -11.314 16.0 150 -13.856 -8.0 16.0 180 -16.0 0.0
16.0 210 -13.856 8.0 16.0 225 -11.314 11.314 16.0 240 -8.0 13.856 16.0 270 0.0 16.0 16.0 300 8.0 13.856 16.0 315 11.314 11.314 16.0 330 13.856 8.0 16.0

As the phase offset a increases beyond 45.degree., a higher percentage of the correlation value begins to come from the Imag*(t) signal path 705 rather than the Real*(t) signal path 704 of the non-coherent CPM correlator 702. At a phase offset of .theta.=90.degree., for example, all correlation is coming from the Imag*(t) signal path 705 and none from the Real*(t) signal path 704. The output 706 of the real CPM correlator 715 and output 707 of the imaginary CPM correlator 715 progress sinusoidally as a function of the phase offset .theta. and can be expressed as:

Thus, maximum correlation of 16 will be achieved regardless of the phase offset .theta.. The use of multiple correlators as configured in the manner shown in FIG. 10 allows successful correlation regardless of which quadrant of FIG. 9 the receiver operates with respect to the transmitter.

It should be noted that at chip times other than multiples of 16Tc (for the example of chip sequence of 16 chips), the correlation output will be a function of the cross correlation value between the i(t.sub.n) and q(t.sub.n) subcodes. The non-coherent CPM correlator of FIG. 10 should perform no worse as far as cross-correlation than a bi-phase correlator with the same code. In other words, if a given code produces a maximum time sidelobe value of 4/16 through bi-phase correlation, then the worst time sidelobe to be expected from the FIG. 10 correlator should also be 4/16.

FIG. 12 is a block diagram of another embodiment of a non-coherent spread spectrum correlator using separable real and imaginary parts of the received spread spectrum signal. The FIG. 12 correlator uses only two shift registers instead of four shift registers and uses only a single power divider having no imposed phase delay for operating on the received signal s*(t) as opposed to three power dividers in the non-coherent correlator illustrated in FIG. 10. The use of a power divider having no imposed phase delay on the received signal s*(t) is an advantage because power dividers which impose a phase delay on the received signal typically operate optimally over only a relatively narrow bandwidth, while the received signal may cover a relatively wide bandwidth.

In FIG. 12, the received signal s*(t) 401 is coupled to a two-register non-coherent CPM correlator 802 for recognizing a chip sequence in the received signal s*(t). The two-register non-coherent CPM correlator 802 comprises a first power divider
803, which produces duplicate signals 804 and 805, each with a 0-degree phase delay. A local oscillator 806 produces a local carrier signal cos .omega..sub.1 t 807, which is coupled to a second power divider 808. The second power divider 808 produces duplicate signals, one signal 809 with a 0-degree phase delay, and another signal 810 with a 90-degree phase delay. The use of the second power divider 808 to generate signals cos .omega..sub.1 and sin .omega..sub.1 from the same local oscillator 806
maintains the phase offset .theta. between .omega..sub.1 and .omega..sub.0 for both cos .omega..sub.1 and sin .omega..sub.1.

The signals 804 and 809 are coupled to a first multiplier 811, which combines its inputs and produces a first product signal 812. The first product signal 812 is coupled to a first low pass filter 813, which produces a first filtered signal 814
which retains its baseband frequency components. The first filtered signal 814 is coupled to a first even-odd correlator 815.

The signals 805 and 810 are similarly coupled to a second multiplier 816, which combines its inputs and produces a second product signal 817. The second product signal 817 is similarly coupled to a second low pass filter 818, which produces a second filtered signal 819 which retains its baseband frequency components. The second filtered signal 819 is similarly coupled to a second even-odd correlator 820.

In the two-register non-coherent CPM correlator 802 depicted in FIG. 12, the Q portion of the Real*(t) signal is the same as the I portion of the Imag*(t) signal, and the Q portion of the Imag*(t) signal is 180-degrees out of phase (i.e., the inverse) of the I portion of the Real*(t) signal. The Q portion of the Real*(t) signal and the I portion of the Imag*(t) signal are stored in and read from the same register 821. Similarly, the Q portion of the Imag*(t) signal and the I portion of the Real*(t) signal are stored in and read from the same register 827. The two-register non-coherent CPM correlator 802 of FIG. 12 operates in a conceptually similar manner to the non-coherent CPM correlator 702 of FIG. 10.

The first even-odd correlator 815 simultaneously recognizes the real i*(t) components and the imaginary q*(t) components, and comprises a register 821 capable of holding a sequence of chips 822. The odd chips are coupled to a real I summer 823, which combines its inputs and produces a real I correlation signal 824. The even chips are coupled to an imaginary Q summer 825, which combines its inputs and produces an imaginary Q correlation signal 826.

The second even-odd correlator 820 simultaneously recognizes the imaginary i*(t) components and the real q*(t) components, and comprises a register 827 capable of holding a sequence of chips 828. The odd chips are coupled to an imaginary I summer 829, which combines its inputs and produces an imaginary I correlation signal 830. The even chips are coupled to a real Q summer 831, which combines its inputs and produces a real Q correlation signal 832.

The real I correlation signal 824 and the real Q correlation signal 832 are coupled to a real summer 833, which combines its inputs and produce a real correlation signal 834. Similarly, the imaginary Q correlation signal 826 and the imaginary I correlation signal 830 are coupled to an imaginary summer 835, which combines its inputs and produces an imaginary correlation signal 836.

The real correlation signal 834 is coupled to a squaring device 837, which computes the square of its input. The imaginary correlation signal 836 is coupled to a squaring device 838, which computes the square of its input. The two squared values are coupled to a summer 839, which combines its inputs and produces a unified correlation signal 840 representing the sum of the squares of the real correlation signal 834 and the imaginary correlation signal 836. The unified correlation signal
840 is coupled to a square root device 841 which takes the square root of its input and generates a final correlation signal 842. The squaring devices 837 and 838, the summer 839, and the square root device 841 collectively compute the root of the sum of the squares of the real and imaginary signals. A Robertson device such as depicted in FIG. 22 and described elsewhere herein may be used to estimate the root of the sum of the squares. The time between separate correlation pulses 843 may be one symbol code time period Ts if CSK is used.

It should be noted that in the non-coherent CPM correlator 702 of FIG. 10 and the two-register non-coherent CPM correlator 802 of FIG. 12, the process of squaring destroys polarity information.

FIG. 13A is a block diagram of a spread spectrum receiver using serial correlation.

The received signal s*(t) 401 is coupled to a coherent serial CPM correlator 902 for recognizing a correlation sequence in the received signal s*(t) 401.

The coherent serial CPM correlator 902 of FIG. 13A comprises a power divider 903, which produces duplicate signals 904 and 905 with a 0-degree phase delay. The signal 904 is coupled to an I multiplier 906. The other input of the I multiplier
906 is coupled to a locally generated signal i(t) cos .omega..sub.0 t that is, the carrier signal combined with the I chip sequence of the correlation sequence. The signal 905 is coupled to a Q multiplier 911, which is coupled to a locally generated signal q(t) sin .omega..sub.0 t, that is, the carrier signal combined with the Q chip sequence of the correlation sequence. The coherent serial CPM correlator of FIG. 13A uses a coherent reference signal having a frequency .omega..sub.0.

The i(t) signal, which is the waveform representing the I chip sequence, and the q(t) signal, which is the waveform representing the Q chip sequence, each comprise tri-valued return to zero (RZ) waveforms, that is, they are +1 to indicate a logical "1", -1 to indicate a logical "0", and 0 to indicate no value, as shown in FIG. 13B. The i(t) signal and the q(t) signal are offset by one chip time from each other in the sense that the i(t) signal has a value of +1 or -1 at each odd chip time but is 0 during the even chip times, and the q(t) signal has a value of +1 or -1 at each even chip time but is 0 during the odd chip is times.

The I multiplier 906 combines its inputs and produces an I product signal 907. The I product signal 907 is filtered by a low pass filter (not shown) and is coupled to an I integrator 908, which integrates its input and dumps the sum under control of a control input 909. The I integrator 908 produces an I correlation signal 910.

The Q multiplier 911 combines its inputs and produces a Q product signal 912. The Q product signal 912 is filtered by a low pass filter (not shown) and coupled to a Q integrator 913, which integrates its input and dumps the sum under control of a control input 914. The Q integrator 913 produces a Q correlation signal 915. Because the i(t) signal and the q(t) signals are tri-valued return to zero waveforms, only one of the integrators 908, 913 changes value at a time.

The I correlation signal 910 and the Q correlation signal 915 are coupled to a summer 916, which combines its inputs and produces a unified correlation signal 917. The unified correlation signal 917 increases progressively in a stepwise fashion and reaches a maximum when full correlation is achieved. If CSK is used, then the largest of the unified correlation signals 917 for a plurality of parallel coherent serial CPM corr